Two terminals quasi resonant tank circuit

ABSTRACT

A power converter includes a transformer, a primary switch, an auxiliary switch, first and second resonance capacitors, and a secondary side rectification means. A switch mode power supply is formed to use reflected voltage and parasitic capacitance as an energy source for a transformer resonance. The auxiliary switch effectively exchanges energy between the primary inductance of the transformer and the first and second resonant capacitors. The auxiliary switch effectively switches the transformer resonance between two distinct frequencies. In one embodiment of the invention, the power converter can be, but is not limited to, a flyback converter and further includes a comparator and a driver. The comparator is for detecting the voltage across the second resonance capacitor and the driver is configured to drive the auxiliary switch based on the output state of the comparator. The resonant nature of the converter provides zero voltage (ZVS) for the primary switch as well as for the auxiliary switch.

RELATED APPLICATIONS

This Application is a divisional application of co-pending U.S. patentapplication Ser. No. 11/706,554, filed on Feb. 14, 2007, and entitled“TWO TERMINALS QUASI RESONANT TANK CIRCUIT,” which claims benefit ofpriority under 35 U.S.C. section 119(e) of the U.S. Provisional PatentApplication 60/773,765, filed on Feb. 14, 2006, and entitled “TWOTERMINALS QUASI RESONANT TANK CIRCUIT,” which are both herebyincorporated by reference.

FIELD OF THE INVENTION

The present invention is generally directed to the field of resonantcircuits. More specifically, the present invention is directed to twoterminals quasi resonant tank circuit.

BACKGROUND OF THE INVENTION

There are several power converter topologies that have been developedover the years, which are intended to improve the power density andswitching efficiency of power converters. An emerging focus of newconverter topologies is to provide a means to reduce or eliminateconverter switching losses, while increasing the switching frequencies.Lower loss and higher switching frequency means more efficientconverters, which can reduce the size and weight of convertercomponents. Additionally, with the introduction of high speed compositesemiconductor switches, such as metal oxide semiconductor field effecttransistor (MOSFET) switches operated by pulse width modulation (PWM),recent forward and flyback topologies are now capable of operation atgreatly increased switching frequencies, such as, for example, up to 1.0MHz.

However, an increase in switching frequency can cause a correspondingincrease in switching and component stress related losses, as well asincreased electromagnetic interference (EMI), noise, and switchingcommutation problems, due to the rapid ON/OFF switching of thesemiconductor switches at high voltage and/or high current levels.Moreover, modern electronic components are expected to perform multiplefunctions, in a small space, efficiently, and with few undesirable sideeffects. For instance, a modern voltage converter that provides forrelatively high power density and high switching frequencies, shouldalso include uncluttered circuit topologies, provide for isolation ofthe output or “load” voltage from the input or “source” voltage, andalso provide for variable step-up or step-down voltage transformation.

In an effort to reduce or eliminate the switching losses and reduce EMInoise the use of “resonant” or “soft” switching techniques has beenincreasingly employed in the art. The application of resonant switchingtechniques to conventional power converter topologies offers manyadvantages for high density, and high frequency, to reduce or eliminateswitching stress and reduce EMI. However, the complexity required toprovide control to the power switches (illustrated below as S1 and S2),and the components associated with complex control, create a limited usein commercial applications.

Conventional Flyback Voltage Converter Topology

FIG. 1 illustrates a flyback type voltage converter 100. The converter100 includes a transformer 102, a resistor 104, two capacitors 106 and112, and two diodes 108 and 110. The resistor 104 and the capacitor 106are coupled in parallel. One node of the parallel resistor 104 and thecapacitor 106 is coupled to a first terminal of the primary winding ofthe transformer 102. An anode of the diode 108 is coupled to the primaryturns of the transformer 102, and the cathode of the diode 108 iscoupled to the other node of the parallel resistor 104 and capacitor106. An input voltage V_(IN) is coupled to a first terminal of theresistor 104 and to a ground terminal. An anode of the diode 110 iscoupled to a first terminal of a secondary winding of the transformer102. A cathode of the diode 110 is coupled to a first terminal of thecapacitor 112. A second terminal of the capacitor 112 is coupled to thesecond terminal of the secondary winding of the transformer 102. A firstterminal of a switching component 115 is coupled to a second terminal ofthe primary winding of the transformer 102 to provide ON and OFF inputpower cycles to the transformer 102. A second terminal of the switchingcomponent 115 is coupled to a sense resistor 117, which in turn iscoupled to ground. A load 114 is typically coupled to the output of theconverter 100, at the secondary turns of the transformer 102.

The Flyback topology has long been attractive because of its relativesimplicity when compared with other topologies used in low powerapplication. The flyback “transformer” serves the dual purpose ofproviding energy storage as well as converter isolation, theoreticallyminimizing the magnetic component count when compared with, for example,forward converter. A drawback to use of the flyback is the relativelyhigh voltage and current stress suffered by the switching components.Additionally, high turn-off voltage (caused by the parasitic oscillationbetween transformer leakage inductance and switch capacitance) seen bythe primary switch traditionally requires the use of a RCD 108,106,104.This parasitic oscillation is extremely rich in harmonics and pollutesthe environment with EMI, and causes high switching losses from theswitching components in the form of extra thermal dissipation. Theseswitching losses are further described below in relation to FIG. 2.

Conventional Flyback Timing Diagram

Accordingly, the converter 100 is configured to receive the inputvoltage V across the primary turns of the transformer 102, and providepower through the secondary turns of the transformer 102, to a loadrepresented by the resistor 114. Also shown in FIG. 1, current at theprimary side of the transformer 102 is proportional to the currentflowing through the sense resistor and is represented by I_(PRI), whilecurrent at the secondary side is shown by I_(SEC).

The flyback voltage converter 100 suffers from loss, noise, and otherinefficient and/or undesirable effects during operation. For instance,FIG. 2 illustrates a diagram 200 of the voltage and current signalcurves recorded during the operation of the flyback converter 100 ofFIG. 1. A shown in FIG. 2, the diagram 200 includes signals for theinput voltage V_(IN), and a drain to source voltage V_(DS,) across theswitch 115 and current I_(PRI) through the switching component 115, attimes t1 through t5. Also shown in this figure, the illustrated signalcurves include noise effects and a saw tooth shape that result from thehard switching of the flyback converter 100. The harsh electronic noiseeffects from ringing are particularly dramatic about the hard switchtimes of the switching cycle. Further, as mentioned above, theseundesirable effects become even more pronounced at the higher switchingfrequencies required by modern voltage converter applications.

In an effort to reduce or eliminate the switching losses and reduce EMInoise caused by high switching frequencies, “resonant” or “soft”switching techniques are increasingly being employed. Resonant switchingtechniques generally include an inductor-capacitor (LC) subcircuit inseries with a semiconductor switch which, when turned ON, creates aresonating subcircuit within the converter. Further, timing the ON/OFFcontrol cycles of the resonant switch to correspond with particularvoltage and current conditions across respective converter componentsduring the switching cycle allows for switching under zero voltageand/or zero current conditions. Zero voltage switching (ZVS) and/or zerocurrent switching (ZCS) inherently reduces or eliminates many frequencyrelated switching losses.

Several power converter topologies have been developed utilizingresonant switching techniques, such as, for example, U.S. Pat. No.5,694,304 entitled “High Efficiency Resonant Switching Converters,” toTelefus, et al., (Telefus), which is hereby incorporated by reference;U.S. Pat. No. 5,057,986 entitled “Zero Voltage Resonant TransitionSwitching Power Converter,” to Henze, et al., (Henze), which is herebyincorporated by reference; U.S. Pat. No. 5,126,931 entitled “FixedFrequency Single Ended Forward Converter Switching at Zero Voltage,” toJitaru (Jitaru), which is hereby incorporated by reference; and U.S.Pat. No. 5,177,675 entitled “Zero Voltage, Zero Current, ResonantConverter,” to Archer, (Archer), which is hereby incorporated byreference.

In particular, Henze describes single ended DC-DC flyback topologies foroperation at very high switching frequencies, such as 1.0 MHz orgreater. In Henze, a plurality of pulse width modulated (PWM) switchesare utilized to effect zero voltage resonant transition switching.Jitaru describes variations of known forward and/or flyback convertertopologies employing zero voltage and/or zero current resonanttechniques. Jitaru specifically describes a forward converter topologyutilizing resonant switching techniques to operate at constantfrequency. Archer describes zero voltage, and zero current, switchingtechniques in resonant flyback topologies utilizing a resonanttransformer assembly inserted in parallel with either the primary orsecondary winding of the main transformer.

The application of such resonant switching techniques to conventionalpower converter topologies offers many advantages for high density, highfrequency converters, such as quasi sinusoidal current waveforms,reduced or eliminated switching stresses on the electrical components ofthe converter, reduced frequency dependent losses, and/or reduced EMI.However, energy losses incurred during control of zero voltage switchingand/or zero current switching, and losses incurred during driving, andcontrolling the resonance means, are still problematic. For instance,some researchers have implemented an active clamp in conjunction with aresonant converter circuit to realize the benefits of high frequencyswitching, while reducing its many side effects. See, for example, theUnited States Patent to Telefus, incorporated by reference above.

SUMMARY OF THE DISCLOSURE

A power converter includes a transformer, a primary switch, an auxiliaryswitch, first and second resonance capacitors and secondary siderectification means. When the auxiliary switch is on, a first resonancefrequency is formed by the energy exchange between the primaryinductance of said transformer and said first resonance capacitor. Whenthe auxiliary switch is off, a second resonance frequency is formed bythe exchange of energy between said transformer and said first andsecond resonance capacitors.

In one embodiment of the invention, the power converter can be, but isnot limited to, a flyback converter and further includes a comparator todetect the voltage across the second resonance capacitor and drivermeans to drive the auxiliary switch based on the output state of saidcomparator.

It is an object of the invention to provide substantially Zero VoltageSwitching (ZVS) for the primary switch. It is a further object of theinvention to provide substantially Zero Voltage Switching for theauxiliary switch. It is another object of the invention to include theenergy stored in most parasitic capacitances in the primary switch, thesecondary switch and the transformer in the resonance cycle. It is yetanother object of the invention to provide driver means for theauxiliary switch that is independent from the driving means for theprimary switch. It is yet another object of the invention to extract theenergy for said driver means for the auxiliary switch from the mainresonance cycle, making the auxiliary switch substantially self driven.

BRIEF DESCRIPTION OF THE DRAWINGS

The novel features of the invention are set forth in the appendedclaims. However, for purpose of explanation, several embodiments of theinvention are set forth in the following figures.

FIG. 1 illustrates a conventional flyback topology.

FIG. 2 illustrates a timing diagram for the flyback topology of FIG. 1.

FIG. 3 illustrates a two terminal resonant tank circuit of the presentinvention.

FIG. 4 illustrates an alternate embodiment of a two terminal resonanttank circuit with parasitic capacitances.

FIG. 5 illustrates the circuit of FIG. 4 with parasitic capacitances andresonances.

FIGS. 6A, 6B, 6C, 6D and 6E illustrate timing diagrams for the operationof the circuits illustrated in FIGS. 3, 4 and 5.

FIG. 7 is a signal diagram having a quasi or sinusoidal shape.

DETAILED DESCRIPTION

In the following description, numerous details and alternatives are setforth for purpose of explanation. However, one of ordinary skill in theart will realize that the invention can be practiced without the use ofthese specific details. In other instances, well-known structures anddevices are shown in block diagram form to not obscure the descriptionof the invention with unnecessary detail.

A detailed description of the principles of operation will be givenbased on the preferred embodiment of the invention in a power converterof the Quasi Resonant Flyback type. The invention can also be used onother converter types such as, but not limited to, a forward converter.

The circuit of FIG. 3 illustrates a conceptual representation of theinvention in a Quasi Resonant Flyback converter. The power converter inFIG. 3 comprises of a transformer (303) with primary and secondarywindings, a primary switch (305), an auxiliary switch (304), a firstresonance capacitor (306), a second resonance capacitor (302) and acomparator (309) with driving means for the auxiliary switch (304). Theconverter further includes secondary rectifier means comprising of adiode (307) and a capacitor (308).

The circuit in FIG. 3 further includes a DC power source (301) toprovide power to the primary side of the power converter. The comparator(309) and driver means for auxiliary switch (304) are configured suchthat when the voltage across the primary winding of the transformer(303) is higher than zero, the auxiliary switch will be in the onposition. The comparator (309) and driver means for auxiliary switch(304) are further configured such that when the voltage across theprimary winding of the transformer (303) is equal or lower than zero,the auxiliary switch will be in the off position. Consequently a firstresonance frequency exists for voltages of less than or equal to zeroacross the primary winding of transformer (303) as a result of theenergy exchange between the primary inductance of transformer (303) andthe first resonance capacitor (306).

For voltages of higher than zero across the primary winding of thetransformer (303), the auxiliary switch (304) will be in the on positionand the second resonance capacitor (302) is connected in parallel withthe first resonance capacitor (306). Consequently, a second resonancefrequency, which is lower in value than said first resonance frequency,exists for voltages of higher than zero across the primary winding oftransformer (303) as a result of the energy exchange between the primaryinductance of transformer (303) and the first and second resonancecapacitors (306 and 302).

FIG. 6A represents the drive signal Vgs1 for the primary switch (305),and the voltage Vds1 across the primary switch (305). The followingdescription of a single switching cycle of the power converter from FIG.3 is based on a steady state continuous waveform under minimum load atthe output of the power converter. The invention also provides similarbenefits when operating under other circumstances, such as, but notlimited to, operation under a load at the output of the power converterand during start-up of the power converter.

When the power converter from FIG. 3 operates under minimum loadconditions, the duty cycle of the drive signal (Vgs1) needs to be verysmall. One switching cycle of the power converter is now discussed fromthe moment that the primary switch (305) is switched off until themoment that said primary switch is switched on again. At switch off ofthe primary switch (305) the voltage across the first resonancecapacitor (306) is substantially equal to the voltage of the powersource (301). As a result of the resonance between the inductance of theprimary winding of the transformer (303) and the first resonancecapacitor (306), the voltage across the primary switch will increase andconsequently the voltage across the first resonance capacitor (306) andthe primary winding of transformer (303) will reduce.

After a quarter cycle of the first resonance frequency, the voltageacross the primary switch (305) will be substantially equal to thevoltage of power source (301) and consequently the voltage across thefirst resonance capacitor (306) and the primary winding of thetransformer (303) is substantially zero. At this moment most of theenergy originally stored in the first resonance capacitor (303) is nowstored in the inductance of the transformer (303). Also at this momentthe auxiliary switch (304) is switched on by comparator (309) and thedriving means for the auxiliary switch. As a result of the secondresonance between the inductance of the primary winding of thetransformer (303) and the first and second resonance capacitors (306 and304) in parallel, the voltage across the primary switch will furtherincrease and consequently the voltage across the first resonancecapacitor (306) and the primary winding of transformer (303) will alsoincrease.

After a quarter cycle of the second resonance frequency, most of theenergy stored in the inductance of the transformer will be transferredto the first and second resonance capacitors but in opposite polarityand lower amplitude (Vreset) compared to the original start voltage(Vsource) across the first resonance capacitor (306). As a result of theresonance between the first and second resonance capacitors and theprimary inductance of the transformer, the voltage across the primarywinding and consequently the voltage across primary switch (305) willstart to decline.

After a quarter cycle of the second resonant frequency, most of theenergy stored in the first and second resonance capacitors will again bestored in the inductance of the transformer. At that moment the voltageacross the first and second resonance capacitors and the primary windingof the transformer is substantially zero. At that moment the auxiliaryswitch will be switched to the off position again by the comparator andthe driver means for the auxiliary switch. After the auxiliary switch isswitched to the off position, the resonance will continue based on theoriginal first resonance between the inductance of the primary windingof the transformer and the first resonance capacitor.

After a quarter cycle of the first resonance frequency, the voltageacross the primary switch (305) will further reduce until it reachessubstantially zero. At that moment the primary switch can switch onunder substantially Zero Voltage Switching conditions. The auxiliaryswitch (302) also switches under substantially Zero Voltage Switchingconditions as a result of the comparator (309) which detects the zerovoltage point across the primary winding of the transformer (303), whichcoincides with a substantially zero voltage across the auxiliary switch.

FIG. 6B represents the drive signal (Vgs1) for the primary switch 305,and the voltage (Vds1) across the primary switch 305 in a situation thatthere is a significant load connected to the secondary side of the powerconverter of FIG. 3. In this situation, the primary switch (305) isswitched on for a longer period in order to charge a larger amount ofenergy into the inductance of the transformer (303). After the primaryswitch 305 is switched to the off position, the rise of the voltage(Vds1) across the primary switch 305 progresses in a similar way asdescribed above for the zero load condition.

Once the voltage across the primary switch 305 reaches the level ofVreset, the additional energy in the transformer 303 caused by thelonger charge period during the on state of the primary switch 305,discharges through the secondary side of the transformer (303) via thesecondary rectifier diode (307) into the secondary smoothing capacitor(308) and eventually into the load (310).

FIG. 6C represents the current (I_(d)) through the secondary rectifier307 relative to the voltage (Vds1) across the primary switch 305.

FIG. 4 represents a more practical realization 400 of the preferredembodiment of the current invention. The auxiliary switch is representedby a MOSFET (420) with its parasitic capacitances Ciss1 (442), Coss1(428) and Crss1 (440), and its inherent body diode (422). The auxiliaryswitch is represented by a MOSFET (424) with its parasitic capacitancesCiss2 (446), Coss2 (448) and Crss2 (444), and its inherent body diode(426). The circuit 400 further includes a transformer (402), drivingcircuitry for the auxiliary MOSFET comprising of three diodes (430, 432,434) and a capacitor (436), and secondary rectification means comprisingof a rectifier diode (410) and a smoothing capacitor (412).

In slightly simplified form, the first resonance capacitor (302) as itappears in FIG. 3 is represented, in FIG. 4, by the addition of Coss1(428) and Crss1 (440) in series with Cres (438) plus the addition ofCoss2 (448) and Crss2 (444). The second resonance capacitor (306), ofFIG. 3, is represented in FIG. 4 by the capacitor Cres (438). As aresult of the diodes in the circuit and the switching of the auxiliaryMOSFET the effective resonance capacitance, and consequently theresonance frequency, will be slightly different for the four quadrantsof the switching cycle in comparison to the resonance frequencies in thesimplified circuit of FIG. 3 as can be calculated by one with ordinaryskill in the art.

In most single ended power converters such as the flyback converters ofFIGS. 3 and 4, it is desirable to keep the reset voltage limited so thatthe voltage level across the switching MOSFET (Vds1) remains within thesafe operating area. In this situation the reset voltage (Vres) acrossthe primary winding of the transformer (402) is lower than (Vde) thevoltage across the primary winding of the transformer during the on-timeof the primary MOSFET (424). To achieve Zero Voltage Switching for theprimary MOSFET the energy (Ehigh) in the effective resonance capacitanceat the point of maximum reset voltage (Vres) has to be equal or largerthan the energy (Elow) in the effective resonance capacitance just priorto the switch on of the primary MOSFET.

The following equations express the values of Elow and Ehigh as afunction of voltages and capacitances:Elow=(Vsource²(Coss1+Coss2+Crss1+Crss2))/2  EQ1Ehigh=(Vres2(Cres+Coss1+Crss1+[Ciss1·Cdv]/[Ciss1+Cdv]))/2  EQ2To meet ZVS for the primary MOSFET the following equation has to be met:Ehigh=Elow  EQ3The value for Cres can be determined from equations EQ1, EQ2 and EQ3.

The preferred embodiment of the invention as represented in FIG. 4further includes a method for driving the gate of the auxiliary MOSFETfrom the voltage across Cres (438). At the moment that the rising edgeof the voltage across the primary winding of the transformer reacheszero, the body diode 422 of the auxiliary MOSFET (420) startsconducting, which effectively switches on the auxiliary switch (304) asit appears in the simplified circuit of FIG. 3. Also diode (432) startsconducting at this point and starts charging Ciss1 (442) through Cdv(436). The further rising voltage across Cres (438) is divided by thegate drive capacitor Cdv (436) and the addition of parasiticcapacitances Ciss1 (442) and Crss1 (440).

When the voltage across Ciss1 reaches the threshold voltage of auxiliaryMOSFET (420), said MOSFET will turn on. It is important that the ratiobetween Vres and the maximum voltage on the gate of MOSFET (420) ischosen to stay within the safe operating area of said MOSFET. Said ratiocan be dimensioned with the value of the driver capacitor Cdv (436). Thegate voltage of MOSFET (420) will remain substantially the same untilthe voltage across Cres has reduced to the same level of said gatevoltage. When the voltage across Cres further reduces, diode (434)starts conducting and will pull the gate voltage of MOSFET (420) downuntil it reaches the gate threshold voltage at which point the auxiliaryMOSFET (420) switches off. Diodes (430, 432 and 434) further prohibitthe voltage across Cres (438) to go significantly below zero.

FIGS. 6C and 6D represent the gate voltage of the auxiliary MOSFET (420)relative to the voltage (Vds1) across the primary MOSFET (424),respectively at zero load (FIG. 6D) and under normal load (FIG. 6E).

In a practical circuit using an embodiment of the invention such as, butnot limited to, the circuit of FIG. 4, the transformer 402 may includeother parasitic components such as, but not limited to, a leakageinductance. An additional resonance between the first and secondresonance capacitors and other parasitic components of the transformermay cause additional voltage fluctuations that are superimposed on thewaveforms as represented in FIGS. 6A through 6E. These additionalvoltage fluctuations may distort the waveforms but do not inhibit theprinciple operation of the current invention.

As mentioned above, the circuit 300 of FIG. 3 further includes certainparasitic capacitances that store and release energy. FIG. 4 illustratesan alternative implementation for the resonant flyback topology of FIG.3, with representative parasitic capacitances, while FIG. 5 shows thecircuit 400 of FIG. 4, with additional resonances. As shown in FIG. 5,the circuit 500 includes nonparasitic (real) and parasitic capacitancesC 503, C_(DV) 536, C_(RES) 538, C_(rss1) 540, C_(iss1) 542, C_(rss2)544, C_(iss2) 546, C_(oss1) 528 and C_(oss2) 548, and switches S1 andS3. The switch S1 of some embodiments serves as a primary switch and isformed by using a MOSFET 524, a diode 526, and a capacitance C_(oss2)548, all coupled in parallel. Accordingly, the gate lead of the MOSFET524 is typically coupled to a controller and/or a driver for the circuit500. The circuit 500 also includes a diode 510, a capacitor 512, and aload 514 coupled to the secondary turns of a transformer 502, whichtypically includes a rectifier type circuit.

FIG. 5 further illustrates that the primary turns of the transformer 502have inductances L 501, L_(RES) 516 and L_(MAG) 518. Hence, thetransformer 502 is coupled in parallel to the capacitor C_(rss2) 544,and selectively coupled in parallel to the capacitor C_(rss1) 540through the switch S2, in an inductor-capacitor (LC) circuit typearrangement. For instance, the capacitor C_(rss1) 540 is selectivelycoupled and de-coupled from the inductor-capacitor arrangement by usingthe switch S2 to advantageously vary the properties of the circuit, suchas to produce multiple frequencies, for example. As shown in theexemplary embodiment 500 of FIG. 5, the switch S2 serves as an auxiliaryswitch and comprises a MOSFET 520, a diode 522, and a capacitanceC_(oss1) 528, all coupled in parallel. The switching frequencies arefurther described below and in relation to FIG. 7.

Zero Voltage Switching

The circuits 300, 400 and 500, of FIGS. 3, 4 and 5, advantageouslyoperate by using two distinct frequencies and zero voltage switching.These two frequencies are further described in relation to FIG. 3. Thefirst frequency f₁ is generated when the switch 304 is activated. Whenthe switch 304 is activated or ON, the capacitor 302 is coupled inparallel to the inductor-capacitor circuit that includes the capacitor306 and the primary turns of the transformer 303. Hence, the firstfrequency f₁ is given by:

f₁=1/[2π√{square root over (L·Σ(C₃₀₂+C₃₀₆))}], where L is the inductanceof the primary turns, C₃₀₂ is the capacitance of the capacitor 302, andC₃₀₆ is the capacitance of the capacitor 306, illustrated in FIG. 3; andf ₂=1/[2π√{square root over (L·C ₃₀₆)}].

FIG. 7 conceptually illustrates these two distinct frequencies f₁ andf₂, in relation to the zero voltage switching of some embodiments infurther detail. More specifically, FIG. 7 is a signal diagram 700 havinga quasi or sinusoidal shape. As shown in this figure, the firstfrequency f₁ is given by the portion of the signal curve above theX-axis, while the second frequency f₂ is given by the portion of thesignal curve below the X-axis. Also illustrated in this figure, theswitching from the first frequency f₁ to the second frequency f₂ isadvantageously by zero voltage switching, while the switching from thesecond frequency f₂ to the first frequency f₁ is also by a zero voltageswitch. As mentioned above, such switching is more efficient andproduces fewer undesirable effects such as, for example, EMI, noise, andharmonics.

ADVANTAGES

In the power conversion industry, one of the most traditionalconventional power supply technologies is flyback voltage conversion.Flyback technology converts DC high voltage or DC low voltage by storingand releasing energy. Typically, flyback type conversion is notpreferred for high power applications but is considered good for low tomedium power conversion, of up to about 100 Watts, for example. Hence,flyback technology is still considered a beneficial topology forparticular applications in the power conversion industry in terms of itssmall implementation size, electrical energy efficiency, andfriendliness to the electrical environment (in terms of noise and/or EMIeffects). However, in modern high frequency and/or high powerapplications, conventional technology such as flyback conversion,experiences several undesirable effects as the product of naturalphenomena.

For instance, Land's law and Maxwell's law states that a frequencygenerated magnetic field depends on the operating frequency, such as,for example, the switching frequency in voltage conversion applications.Accordingly, as frequency increases, elements introduced by magneticfields become smaller. Researchers have exploited this property in asolution for the problem of magnetic effects during voltage conversion.More specifically, by increasing switching frequency, the magneticfields and properties naturally generated by the fast switchingelectrical components, become smaller. However, high frequency operationhas tradeoffs. For instance, in addition to increased switching lossesand noise levels, higher frequency operation by using a large amount offast, hard switching, further undesirably creates waveforms withincreasing harmonics. Hence, the high frequency solutions in the artreduce only one undesirable element. Here, only undesirable magneticeffects are typically reduced, while many other undesirable effects aregenerated instead.

These further undesirable tradeoff effects require a panoply ofpatchwork fixes including heat sinks, and larger sized power supplies,that are less efficient and more costly. Thus, the tradeoff of highfrequency for lower magnetics creates a net loss in the system. However,due to the laws of physics, reducing undesirable magnetic effects,mandates the use of higher operating frequencies, and its set ofundesirable drawbacks.

More specifically, in a power supply, the most complicated componentsare typically the transformer(s) and the transistor(s). Increasing theswitching frequency, reduces the cross sectional area of magnetic fieldgeometry of these electrical components, which reduces the undesirablemagnetic effects of the switching. However, the higher operatingfrequency traditionally causes saw tooth type waveforms. These hardswitching waveforms are also indicative of high amounts of noise andlosses due to inefficient switching. Further, these noisy, sawtooth typeof waveforms are rich with harmonics, which result in furtherundesirable effects and inefficiency.

In view of the foregoing, some embodiments employ high frequencyoperation advantageously, by introducing resonant high frequencies,which reduces the effects of hard switching and makes switching“softer.” These embodiments, rather than apply patchwork fixes, changethe fundamental voltage converter circuit, by applying a resonance(frequency). The embodiments described above illustrate the operation ofsuch a resonant type circuit, which make the switched transistor anddiode components of such circuits operate in a ‘soft mode,’ which moreclosely resembles a sine wave. Moreover, these embodiments have no, orminimal, switching losses, because the voltage and/or current approach azero value before the periodic, sinusoidal switch to the rising orfalling portion of the sine wave.

In some embodiments, a capacitor and an inductor form the resonancecircuit, while some embodiments couple a capacitor (a real component) inparallel with the coil of a transformer (an inherent inductor) to form aparallel resonance circuit. In both these types of implementations, theparasitic capacitances advantageously generate a negative current backto the source. Hence, the parasitic capacitances, which are normallyproblematic for a system, operate as a large capacitor that stores andreleases energy in conjunction with the resonant tank and the realcapacitances. Stated differently, all the components are in a fullresonance circuit, including the real or active components, and theparasitic components. Therefore, all or most of the energy generated bythe components of the system, including the parasitic components, istransferred either to the load (typically in the low frequency mode), orback to the source (in the high frequency mode).

In particular, the cyclical operation of some embodiments implement aquasi resonant storage tank by the generation of a series of dampeningsinusoidal wave forms. Some embodiments further maintain, for thesystem, an energy equilibrium such that the energy of a high frequencycycle is substantially equivalent to the energy of a low frequencycycle.

Also, as mentioned above, some embodiments of the invention have onlytwo terminal nodes. These two terminal implementations have benefits interms of both manufacturing and commercial aspects. For instance, theresonant tank circuit of some embodiments has active components thatcouple across the transformer by using only two terminal nodes, providean elegant design for manufacture. Moreover, the two terminal nodedesign results in only two pins, which has certain commercial and/orpackaging advantages.

While the invention has been described with reference to numerousspecific details, one of ordinary skill in the art will recognize thatthe invention can be embodied in other specific forms without departingfrom the spirit of the invention. For instance, in a particularembodiment the drain voltage V_(D) on the primary switch 305 illustratedin FIG. 3 can be on the order of 100 volts, while the gate voltage V_(G)is about 20 volts, the current I_(PRI) is about 200 milli-amperes, andI_(SEC) is about 2.0 amperes, while the inductor of some embodiments maybe rated at about 600 micro-Henry. However, one of ordinary skillrecognizes that these particular signals and values vary for eachspecific power implementation. Thus, one of ordinary skill in the artwill understand that the invention is not to be limited by the foregoingillustrative details, but rather is to be defined by the appendedclaims.

1. A quasi resonant type circuit comprising: a. an inductor; b. acapacitor coupled to the inductor in a parallel inductor-capacitor (LC)type arrangement; c. a transistor coupled to the inductor-capacitor typearrangement, the transistor for providing a switching cycle, wherein theswitching cycle is determined with respect to a charge on the capacitor;d. a quasi resonant storage tank for storing an electromagnetic energyby using a resonant current; wherein the circuit is configured toreceive the resonant current in a first cycle of operation, wherein thecircuit is configured to release the stored energy in a second cycle ofoperation.
 2. The circuit of claim 1, wherein the circuit has anassociated signal curve that is sinusoidal.
 3. The circuit of claim 1,wherein the circuit is configured for operation by using the first andsecond cycles without the need for an external driver.
 4. The circuit ofclaim 1, wherein the circuit is configured for operation by using thefirst and second cycles without the need for external control.
 5. Thecircuit of claim 1, further comprising: a set of parasitic components;and a set of non parasitic components, wherein the parasitic and the nonparasitic components store the energy, wherein the energy is retrievedfrom the parasitic components.
 6. The circuit of claim 1, wherein thetransistor comprises a MOSFET.
 7. The circuit of claim 1, wherein aninput power across the primary turns of the transformer provides powerto the circuit.
 8. The circuit of claim 1, the circuit consisting of nomore than two terminal nodes within the circuit, wherein the twoterminal nodes are coupled to no more than two pins.
 9. The circuit ofclaim 1, wherein the circuit is coupled to a transformer for a powerconverter, wherein the circuit provides the switching cycle for thepower converter.
 10. The circuit of claim 9, wherein the power convertercomprises a flyback type power converter.
 11. The circuit of claim 9,wherein the power converter comprises a forward type power converter.12. The circuit of claim 1, wherein the transistor is coupled in aparallel arrangement to the LC type arrangement.
 13. The circuit ofclaim 1, wherein the first and second cycles of operation correspond tothe switching cycle.